Signal transmission system

ABSTRACT

Systems and methods of signal transmission and measuring for sensors employing a transmission medium are provided. In one embodiment, a method may comprise measuring a first monitored condition to generate a first monitored condition signal; converting the first monitored condition signal to a first frequency modulated signal having a first frequency; generating a second frequency modulated signal having a reference frequency; transmitting the first frequency modulated signal and the second frequency modulated signal using time division multiplexing; and wherein a first ratio of the first frequency and the reference frequency is associated with the first pressure.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation claiming priority under 35 U.S.C.§120 to U.S. patent application Ser. No. 13/252,840, filed Oct. 4, 2011,which is a continuation claiming priority to U.S. patent applicationSer. No. 12/750,173, filed Mar. 30, 2010, now U.S. Pat. No. 8,122,770,issued Mar. 28, 2012, which is a continuation claiming priority to U.S.patent application Ser. No. 12/228,399, filed Aug. 11, 2008, now U.S.Pat. No. 7,685,880, issued Mar. 30, 2010, which is a continuationapplication claiming priority to U.S. patent application Ser. No.11/803,128, filed May 11, 2007, now U.S. Pat. No. 7,409,866, issued Aug.12, 2008, all of which are entitled “SIGNAL TRANSMISSION SYSTEM,” andall of which are incorporated by reference in their entirety as if fullyset forth herein.

FIELD OF THE INVENTION

This invention relates to a signal transmission system for sensors andmore particularly to a signal transmission and measuring system forsensors employing a single transmission wire and a grounded return.

BACKGROUND OF THE INVENTION

As one can ascertain, the prior art is replete with pressure transducersor sensors employed in harsh environments. Such environments includedeleterious substances which may destroy the transducer, as well as highpressures and temperatures. High temperatures include those temperatureswhich are found in various high temperature environments as combustionengines, for example. In other applications, such as the use of pressuretransducers in injection molding and for other environments extremelyhigh temperatures are also found. The prior art has disclosed pressuretransducers which are capable of operating at very high temperatures astemperatures in excess of six hundred degrees Celsius (600° C.). See forexample, U.S. Pat. No. 7,124,639, which issued on Oct. 24, 2006,entitled “ULTRA HIGH TEMPERATURE HERMETICALLY PROTECTED WIREBONDEDPIEZORESISTIVE TRANSDUCER,” by A. D. Kurtz et al. and assigned to KuliteSemiconductor Products, Inc., the assignee herein. See also U.S. Pat.No. 6,363,792, entitled “ULTRA HIGH TEMPERATURE TRANSDUCER STRUCTURE,”issued on Apr. 2, 2002 to A. D. Kurtz et al. and assigned to theassignee herein. See also U.S. Pat. No. 6,530,282, entitled “ULTRA HIGHTEMPERATURE TRANSDUCER STRUCTURE,” issued on Mar. 11, 2003 to A. D.Kurtz et al. and assigned to Kulite Semiconductor Products, Inc., theassignee herein.

By referring to the above noted patents, one can see applications ofsuch transducers in high temperature environments as well as themonitoring of such signals in such environments. One problem is foundwhen one deals in the oil and geothermal exploration fields. In such oiland geothermal explorations, one uses pressure or temperaturetransducers which are exposed to temperatures much higher than thoseexperienced by standard electronics. Typical transducers which are usedfor normal operations are usually limited to temperatures below onehundred and twenty-five degrees Celsius (125° C.). Due to the depth ofdrilling as well as the use of steam to extract the oil the operatingtemperature in such explorations exceed two hundred degrees Celsius(200° C.). Pressure transducers using a piezoresistivesilicon-on-insulator (SOI) structure are widely used in suchapplications. Such transducers for example are described in the abovenoted patents. Also used are platinum resistors (RTD) used to measurethe temperature which resistors are also capable of operating at thesehigh temperatures. Thus, the combination afforded in regard to the aboveis that one requires a pressure transducer which can operate at hightemperatures and one also requires electronics which can operate at suchtemperatures. See for example a co-pending application entitled “HIGHTEMPERATURE PRESSURE SENSING SYSTEM,” U.S. patent application Ser. No.11/234,724, filed on Sep. 23, 2005 for A. D. Kurtz et al. and isassigned to the assignee herein. In that application, there is describeda high temperature pressure sensing system which includes a transducerhaving pressure sensing piezoresistive elements formed by a SOI process.The system also uses SOI CMOS electronic circuitry which is operativelycoupled to the piezoresistive sensor as well as ancillary circuitryconnected to the unit to provide compensation and normalization. Thatapplication is incorporated by reference in its entirety herein.

Other examples of SOI technology may be seen in U.S. Pat. No. 5,955,771,entitled “SENSOR FOR USE IN HIGH VIBRATIONAL APPLICATIONS AND METHODS OFFABRICATING THE SAME,” issued to A. D. Kurtz and U.S. Pat. No.4,672,354.

In existing oil and geothermal applications, due to the depth of thedrilling as well as due to the use of steam to extract the oil, veryhigh temperatures are involved. In oil and geothermal explorations thewires used in these systems are extremely long and can be as long as tenthousand (10,000) meters. These wires apart from being extremely longare also expensive. The cost of the wire often exceeds the cost of thetransducers. In prior art applications, the pressure transducers areconnected to the wiring via a four-to-twenty milliamp electronicinterface. The second wire is the metal conduit in which the wire isinserted. The prior art method has significant temperature limitationswhich are further aggravated by the significant power dissipation of thefour-to-twenty milliamp interface. This power dissipation increases thejunction temperature of the electronics by several tens of degrees aboveambient temperature. The prior art method also requires a separate wirefor each pressure or temperature sensor.

The present invention discloses a way of interfacing one or morepressure sensors to a measuring system using only one wire for thesignal and power and a return wire which is usually the conduit of thesignal/power wire. An electronic interface is advantageous for sensorslocated in a very high temperature environment at great distances fromthe measuring system such as described above in the oil and geothermalexplorations. The invention is also well suited for integration in acircuit using technology suitable for high temperature operation as thesilicon-on-insulator (SOI) process. The signal transmission system orwire interface described is also applicable and advantageous for use insystems operating at more benign temperatures and over shorter distancesas it simplifies the wiring as well as the measuring method.

SUMMARY OF THE INVENTION

Apparatus for transmitting a transducer signal to be measured from asignal generation location to a measuring location connected by a singlewire where undesirably the transducer signal is subjected to variationscaused by multiple sources. The apparatus comprises a transducerpositioned at the signal generation location and operative when biasedby a power source to provide an output signal according to a monitoredcondition. A reference level generator is coupled to the power sourceand operative to provide a reference level output proportional to thevalue of the power source. A multiplexer for receiving at one multiplexinput the transducer output signal and at another input the referencelevel output to provide at a multiplexer output the transducer signalfor a first interval and the reference level for a second interval. Aconverter responsive to the multiplexer output for converting thetransducer signal to a first frequency modulated signal having afrequency output variation according to the value of the transduceroutput signal during the first interval and for providing a secondfrequency modulated signal indicative of the reference level during thesecond interval, where any variations in signals which may be caused bymultiple sources are present in both signals; and measuring arrangementpositioned at the measuring location and responsive to the modulatedsignals to provide the ratio of the periods of the signals, where theratio is a direct measure of the transducer output signal with theundesired variations substantially eliminated.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 is a block diagram of a one wire system directed from a signalgeneration located to a measuring location according to an embodiment ofthe invention.

FIG. 2 is a block diagram of a wire interface located at the signalgeneration location according to an embodiment of the invention.

FIG. 3 consists of FIGS. 3A and 3B and depict timing diagrams as showingthe pressure and reference intervals according to an embodiment of theinvention.

FIG. 4 consists of timing diagrams showing the output of a capacitor anda monostable multivibrator operating according to an embodiment of theinvention.

FIG. 5 is a block diagram of a signal measuring arrangement according toan embodiment of the invention.

FIG. 6 is an alternate embodiment of a signal measuring system accordingto an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Referring to FIG. 1 there is shown a block diagram of a one wire systemfor measuring a transducer output according to this invention. As partof the signal generation location there is a transducer system 10 whichincludes a bridge 11 and associated circuitry 12. The bridge 11 is aWheatstone bridge, which basically is implemented and fabricated by theuse of piezoresistive SOI pressure transducers. The bridge 11 andassociated circuitry 12 are located at the signal generation location.This location may be the bottom of a drilled shaft for oil explorationor for other purposes. Pressure sensors suitable for use in bridge 11are well known, for example may be those and are described in the abovenoted patents and applications. The output of the bridge or pressuretransducer can be compensated using its inherent resistance versustemperature characteristics to provide a stable ratio metric output overa wide temperature range. Transducers which operate accordingly are alsowell known in the art and are described for example also in the abovenoted patents. Coupled to the output of the transducer is an electronicinterface 12. The electronic interface may be fabricated and implementedby SOI electronic circuits including CMOS transistors.

The entire unit 10 as shown in FIG. 1 which consists of the bridge 11 aswell as electronics 12 may be positioned or inserted into a drilledshaft which would be implemented by techniques as for example used inoil well exploration. This shaft for example may be thousands of meterslong. In any event, the entire signal generation apparatus 10 as shownenclosed in the dashed box is positioned near the bottom end of theshaft. The signal generation apparatus has an output 17 which is coupledto a wire 15. The wire 15 runs from the transducer assembly 10 and iscoupled to output 17 to a measuring site or location including ameasuring circuit 18. The length of the wire 15 may be ten thousand(10,000) meters or longer. Also shown is a return conduit 16 which maybe a shield for wire 15 or may be an actual metal or other conduit usedto surround and protect the wire. As seen at the measuring locationwhich is the other end of the shaft and can be a field office or otherground location at normal ambient temperature is the measuring circuit18. The wire 15 is connected to a voltage source VMS (voltage atmeasuring source) via a resistor 19. In the exemplary configuration,shown in FIG. 1 VMS is five (5) volts while resistor 10 is one hundred(100) ohms. The values are by way of example only and other values canbe employed. The VMS source supplies operating potential (+Vcc) to thetransducer system 10 at the signal generation location. This VMS sourceis the sole power source used to bias the bridge 11 as well as tooperate the circuitry 12. The resistor 19 is DC connected to resistor 39(FIG. 2) associated with MOSFET 38 (FIG. 2). In FIG. 1 resistor 39 isshown, by way of example to be nine hundred (900) ohms. Thus, as seen inFIG. 1, the structure 10 which consists of the bridge 11 and theelectronics 12 is located in a high temperature environment such as thatfound at the bottom of a shaft or hole drilled for oil or geothermalexploration and to measure pressure. The output 17 of the system 10 isdirected to a single wire 15 which also is associated with a returnshield or conduit 16. The wire 15 can be more than ten thousand (10,000)meters long and is directed to the monitoring or measuring stationwhereby the output on wire 15 is measured to develop a voltage or anindication at measuring circuit 18 indicative of the pressure or othermonitored condition. While the above noted system shown in FIG. 1depicts the measurement of pressure, it is understood that othermeasurements can be made such as temperature, etc., utilizing the singlewire interface as described.

FIG. 2 shows a block diagram of the one wire interface showing thecircuit details utilized to implement the signal generation system 10 ofFIG. 1. Essentially before proceeding with a detailed discussion of theinterface, a brief description of the operation will be given.

Referring to FIG. 1 the one wire interface transmits the transduceroutput as a frequency modulated signal at output 17. This frequencymodulated signal propagates over wire 15 which also provides the powersupply to the sensor and interface. Also transmitted over the wire 15 isa reference signal which reference signal is processed through the samechain as the transducer output signal. The reference signal is also afrequency modulated signal. The two signals are multiplexed in a timedivision mode, for example one second of the transducer signal followedby one second of the reference signal. The frequencies of the twosignals are measured by the ground equipment as measuring circuit 18 andthen the ratio of the two periods which are the reciprocals of the twofrequencies is calculated. The ratio of the signals is a direct measureof the transducer output, and eliminates all sources of errors. Sucherrors can be significant, due to the high operating temperatures andare caused by multiple sources, as noise source, RF interference and thelike. Therefore, it becomes very difficult to compensate for suchsources. As the transducer output and the reference signal are passedthrough the same chain, and are affected by the same errors, the ratiocalculation eliminates all errors. It does not matter how these errorsare derived or generated, as they will be present both on the transduceroutput signal as well as on the reference signal and therefore can beeliminated by the apparatus and methods depicted herein.

As one will understand, the technique and apparatus can be furtherenhanced such that multiple sensors can be processed through the samechain and the data sent as time multiplexed signals followed by thereference signal. In this way several sensors can be connected to theground measuring system 18 via a single wire 15. It is also understoodthat temperature sensors can also be used with this interface. A RTDelement can be mounted in a bridge configuration using three fixed metalfilm resistors and the output of the bridge multiplexed and sent throughthe same chain as the reference signal. The reference signal should bevery consistent and stable with temperature in order to enhanceoperational effectiveness.

Referring to FIG. 2 the reference signal is derived from the powersupply +Vcc by a resistive divider which is located on the SOI chip.This implementation has shown that the ratio of resistors on the chip isvery stable and consistent from chip to chip. This ratio is determinedby the geometric features of the resistors and stays constant eventhough the value of the individual resistors may change over temperatureand from device to device. Thus the techniques and apparatus describedherein, also eliminates the errors due to variations in the supplyvoltage. Such variations are possible and expected due to the resistanceof extremely long wires. As the transducer output and the resistivedivider output which is the reference signal are proportional to thesupply voltage +Vcc, the ratio calculation eliminates the undesirederror as well. As one will understand the electronic interface as forexample the circuitry 12 of FIG. 1 is integrated on a SOI chip and theoperating temperature of the interface exceeds two hundred and fiftydegrees Celsius (250° C.), for example. As one can also understand, byreferring to the above noted co-pending application entitled “HighTemperature Pressure Sensor System” the SOI circuitry depicted thereincan be employed herein as well, as for example, FETs, counters, and thelike.

Referring to FIG. 2, there is a shown a circuit diagram in block form ofthe electronic interface 10. As seen the transducer 20 is arranged as aWheatstone bridge configuration. The Wheatstone bridge includes fourpiezoresistors such as 21, which are wired in a bridge configuration.The bridge has a ratiometric output which is compensated over thetemperature range. As seen the bridge derives its biasing voltage fromthe voltage source +Vcc which is applied to the bridge via a spanresistor 22. Thus the bridge 20 produces a ratio metric output which iscompensated over the entire temperature range of operation. Such bridgecircuits including those having ratiometric outputs are well known inthe prior art and examples of such bridge circuits employingpiezoresistors are indicated in the above cited patents. As seen theoutput of the bridge is applied to the input terminals of input (IN1) ofa multiplexer 26. Also shown is a resistive divider consisting ofresistors 23, 24 and 25. The resistors 23, 24 and 25 are in series withone terminal of resistor 25 coupled to reference potential or ground andone terminal of resistor 23 coupled to the biasing voltage source +Vcc.

It is noted that the biasing source +Vcc for the resistive divider isthe same biasing source employed for the bridge. The junction betweenresistors 23 and 24 is applied to one input terminal of the multiplexer26 (INΦ) while the junction between resistors 24 and 25 is applied tothe other terminal of the multiplexer input (INΦ). It is also notedbefore proceeding further that the resistors 23, 24 and 25 are alsodesignated as R1, R2 and R3. The resistors have been so designated astheir values are used in the mathematics which are pertinent to theoperation of the system. The output of the multiplexer 26 (OUT) isapplied to inputs of an instrumentation amplifier 27. The output of theinstrumentation amplifier 27 is applied to the non-inverting input (+)of an operational amplifier 28. The operational amplifier 28 has afeedback resistor 30 also designated as RF which is coupled to a gaincontrol resistor 29 indicated as R gain. The feedback resistor 30 isconnected and at one terminal to the output of operational amplifier 28and at the other terminal to the inverting input (−) of the operationalamplifier 28. Resistor 29 is coupled between the inverting input (−) ofthe operational amplifier 28 and reference potential. The total gain ofthe amplifier arrangement consisting of instrument amplifier 27 and theoperational amplifier 28 is set by the resistor 29. It is of courseunderstood that the instrumentation amplifier 27 which is also anoperational amplifier has a fixed gain such as a gain of ten (10) forexample. The gain of the operational amplifier is controlled by thevalue of resistor 29 which is well known.

The output of the amplifier designated as V_(X) is coupled to thenegative terminal of comparator 31. The positive input terminal of thecomparator is driven or coupled to a capacitor 37. The capacitor 37 ischarged by a current source 36 which is positioned in series with thesource electrode of the FET 35, also designated as Q1. As seen thecurrent source 36 is also coupled to the +Vcc supply. The FET 35 alsohas its source electrode coupled to the positive input of comparator 31and of course coupled to the non-ground terminal of capacitor 37. Theoutput of the comparator 31 is coupled to the input of a monostablemultivibrator 33 whose output is coupled to the gate electrode of theFET 35.

The output of the monostable is also coupled to a four plus eleven-bit(4+11-bit) counter and logic circuit 34. The output of the logic circuit34 is coupled to the selects input (S) of the multiplexer 26. The outputof the counter 34 is also coupled to the gate electrode of the FET 38having a source electrode coupled to a load resistor 39. The resistor 39is coupled to wire 15 which at the other end has one terminal ofresistor 19 coupled at the measuring end. The other terminal of resistor19 is coupled to the VMS source. The junction between resistor 19(FIG. 1) and resistor 39 is the +Vcc which is the biasing potentialshown in FIG. 2 and used to bias all circuitry as amplifier 27, 28,comparator 31, mono 33, counter 34 and so on.

Operation of the circuit is as follows. The positive input of thecomparator 31 as indicated is driven by the capacitor 37. The capacitor37 is charged by the current source 36 and discharged by the MOSFETtransistor 35. The output of the comparator when present triggers themonostable circuit 33 when the capacitor 37 voltage reaches thepredetermined value designated as V_(X). As one can see the output ofoperational amplifier 28 is V_(X). When the value of capacitor 37 ischarged to V_(X) the comparator 31 produces an output which triggersmonostable circuit 33. The time period of the monostable multivibrator33 is chosen to be as short as possible but long enough to safelydischarge the capacitor 37. In one particular example, a time durationof one microsecond for the output of the monostable 33 is appropriate.The value of the capacitor 37 and of the current source 36 is chosensuch that they can be easily implemented on a SOI chip. The prescalersize included in module 34 is chosen such that the output pulses willhave a sufficiently low frequency to provide a useful signal afterpassing through the very high capacitance and resistance of the verylong single wire 15 connection to the measuring circuit 18. The outputof the monostable 33 as seen is applied to a fifteen-bit (15-bit)counter 34 (4+11 bits). The counter controls the select pin of themultiplexer 26 and also controls the gate of the switch MOSFETtransistor 38. The first four bits of the counter 34 are used as theprescaler and the following eleven (11) bits are used as acounter/sequencer.

The monostable output pulses are first divided by 16 and then the outputis applied to the eleven bit sequencer. Thus, as indicated the countercontrols the select pin of the multiplexer 26 as well as the gate of theMOSFET transistor 38. When MOSFET transistor 38 is turned on the loadresistor 39 is inserted in the circuit increasing the currentconsumption of the circuit. The increase in the current consumptionresults in a voltage drop of about one half volt across the one hundred(100) ohm resistor shown in FIG. 1 and coupled to the input of themeasuring circuit 18. The eleven bit counter 34 controls the select pinof the multiplexer such that for one thousand and twenty-four (1,024)periods or the first interval of the sequencer the transducer output isprocessed by the interface through the inputs IN-1 of the multiplexer26. After this period or interval the reference voltage of fiftymillivolts (50 mV), which is derived from the voltage divider consistingof resistors 23, 24 and 25 is processed through inputs INΦ for anotherpulse period of one thousand and twenty-four (1,024) pulses of theprescaler. This is the reference interval. The output of the prescalerdrives the gate of the MOSFET transistor 38 generating through theon/off switching of the load resistor 39, the square wave currentpulses, which appear as the voltage pulses at the input of the measuringsystem.

Thus, as seen in FIG. 3A, during the first interval A, one thousand andtwenty-four (1,024) measurement pulses are generated producing a firstfrequency modulated signal indicative of the value of the transduceroutput signal. During the next or second interval B, nine hundred andsixty (960) reference pulses are generated producing a second frequencymodulated signal indicative of the reference level output. Thereafter,for a third interval C equivalent to a sixty-four (64) pulse period.Transistor 38 is disabled and no pulses are provided. This sixty-four(64) pulse interval informs the measuring system that the next sequenceof pulses one thousand and twenty-four (1,024) is the transducersequence. After counting one thousand and twenty-four (1,024) pulses thenine hundred and sixty (960) reference pulse interval begins and so on.It is also understood that the sixty-four (64) pulse period could bepositioned between the one thousand and twenty-four (1,024) interval andthe nine hundred and sixty (960) interval and serve the same purpose.These intervals are shown in the timing diagram of FIG. 3A, thesixty-four (64) pulses are intended for allowing the measuring system todiscriminate between the transducer signal phase and the referencesignal phase. FIG. 3B is an expanded time scale showing transducerpulses T_(P) and the frequency variation as well as the reference pulsesT_(R).

The voltage across capacitor 37 and the monostable output are shown inFIG. 4. Assuming that the capacitor 37 is discharged and a transistor 35is off, the capacitor 37 is then charged linearly by the current source36. The capacitor charges until it reaches the voltage V_(X). At thismoment the comparator 31 output changes state triggering the monostable33 for a short period (e.g., about one microsecond). During this timethe capacitor 37 is fully discharged. Afterwards the monostable 33 turnsoff, the transistor 35 is turned off and the cycle repeats. It isunderstood that the time scale shown in FIGS. 3 and 4 is distorted forclarity purposes. The duration of the monostable pulses and thedischarge time of the capacitor are less than that of a fraction of onepercent (1%) of the charging time. FIG. 3B shows the pulses as depictedin FIG. 3A expanded in time. The pulse edges are not very fast, and infact are slowed significantly by the very high capacitance of the longwire. Thus, as seen the pressure pulse designated at T_(P) has arelatively slow rise time and fall time as does the reference pulsesdesigned as T_(R). FIG. 3B as indicated shows an expanded version of thepressure transducer pulse values as well as the reference pulse valuesdepicted in FIG. 3A.

Referring again to FIG. 4, there is shown the time diagram of thecapacitor 37 voltage in the top diagram and the output of the monostablein the bottom diagram. Thus, as seen when the capacitor voltage reachesV_(X) the monostable multivibrator triggers for a duration of onemicrosecond. After the monostable pulse the cycle repeats again asdepicted in FIG. 4.

Additional pressure transducers can be employed and for any additionaltransducer the multiplexer 26 will need an additional set of inputs andthe counter/sequencer circuit 34 is implemented to provide additionalintervals for the second, third and fourth transducer. As can be seen byreferring to FIG. 3A one can implement multiple cycles concerning acycle A, A₁, A₂ followed by a reference cycle B. In the exemplaryconfiguration, the value of resistor 19 of one hundred (100) ohms at theinput of the measuring circuit is arbitrarily chosen. In a preferredembodiment, the value would be equal to the characteristic impedance ofthe wire and the conduit. In this case, the bandwidth of the signaltransmission is significantly higher resulting in a much shortermeasurement cycle than shown allowing multiple transducer data to besent in a shorter time. Also, the pulses shown in FIGS. 3A and 3B willhave much faster transition times resulting in a better accuracy andnoise immunity of the period measurement. Another enhancement of theinterface can be the addition of a thermal electric cooler for theelectronic chip. This chip has a very small size and consequently a verysmall thermal mass. Thus, a small thermal electric cooler could bepositioned on the chip to maintain the chip temperature at safe lowlevels without significant power consumption. Such thermal electriccoolers also designated as PELTIER coolers are well known and areemployed in many electronic chips such as microprocessors for use incomputers and this will not be described in further detail.

In order to more clearly understand the nature of the invention thefollowing circuit analysis is hereby presented.

Referring again to FIG. 2, in conjunction with FIGS. 3 and 4, assumingthe sensor is a piezoresistive bridge 20 compensated using traditionalways, the bridge output voltage V_(BR) can be written as:V _(BR) =k*p*V _(CC),

where k is the bridge sensitivity, p is the pressure, and V_(CC) is thebridge supply voltage. During the measuring phase, i.e. when IN1 of themulitplexer 26 is selected, the output of the amplifier 28 V_(X) can beexpressed as;V _(X) =G*V _(BR),

where G is the amplifier gain.

The voltage u_(c) across the capacitor 37 is:

${u_{c} = \frac{I_{Q}*t}{C}},$

where I_(Q) is the capacitor charging current, t is the time and C isthe capacitance 37. I_(Q) is generated by the current source 36. Whenu_(c) reaches the level V_(X) then the comparator 31 changes state, thustriggering the monostable circuit 33 which rapidly discharges thecapacitor 37 through the transistor 35 (Q1). Neglecting the very shortdischarge time, the cycle time T of the capacitor 37 (C) can becalculated by substituting V_(X) instead of u_(c), resulting:

$T = {\frac{C*V_{X}}{I_{Q}}.}$

Taking into account the prescaler 34 factor of sixteen (16), the periodT_(P) of the output signal during the measuring phase is:

$T_{P} = {\frac{16*C*V_{X}}{I_{Q}}.}$

By substituting the formulas for V_(X) and V_(BR) the period T_(P)becomes:

$T_{P} = {\frac{16*C*G*k*p*V_{CC}}{I_{Q}} = {k*p*{\frac{16*C*G*V_{CC}}{I_{Q}}.}}}$

Considering now the second phase, when the reference voltage V_(R) isselected by the multiplexer 26 we have:

$V_{R} = {{\frac{R\; 2}{{R\; 1} + {R\; 2} + {R\; 3}}*V_{CC}} = {r*{V_{CC}.}}}$

The factor r is the resistance ratio:

$r = {\frac{R\; 2}{{R\; 1} + {R\; 2} + {R\; 3}}.}$

A similar calculation for the period T_(R) during this reference phaseresults:

$T_{R} = {r*{\frac{16*C*G*V_{CC}}{I_{Q}}.}}$

At this point, it is important to note that the capacitance C 37, thegain G, the supply voltage V_(CC) and the charging current I_(Q) allhave large random shifts with temperature and variations from device todevice, while the factor r is very constant over temperature and fromdevice to device. Calculating the ratio between T_(P) and T_(R) results:

$\frac{T_{P}}{T_{R}} = {\frac{k}{r}*{p.}}$

This allows one to calculate the pressure p as:

$p = {\frac{r}{k}*{\frac{T_{P}}{T_{R}}.}}$

This formula shows that the pressure p can be calculated from two timemeasurements of T_(P) and T_(R) and from two very stable and welldefined constants k and r.

For the component values shown on the schematic:

I_(Q)=1 μA,

C=10 pF,

V_(X)=1V to 4V during the measuring phase,

V_(X)=2.5V during the reference phase,

resulting in the charging times of the capacitor to be one hundredmicroseconds (100 μs) to four hundred microseconds (400 μs) during themeasuring phase and two hundred and fifty microseconds (250 μs) duringthe reference phase.

Due to the four-bit (4-bit) prescaler, i.e. divide by sixteen (16), therespective periods in the transmitted waveform are sixteen (16) timeslonger, i.e. T_(P)=1.6 ms to 6.4 ms during the measurement phase, andT_(R)=3.2 ms nominal during the reference phase. The duration of theflat portion of the waveform is sixty-four (64) periods of the referencepulses, corresponding to 204.8 msec.

It is important to note that the same calculations can be done for othermeasurements other than the pressure p. To measure the temperature, abridge with three fixed resistors and an RTD as the fourth arm can beused.

As one can understand the interface as described above has extremeadvantages over the prior art. One main advantage is that there is areduced number wires for multiple transducers resulting in greater costsavings, less complexity, and improved reliability. The electroniccircuit is implemented in a single integrated chip as all the componentsshown in the above diagram of FIG. 2 are known building blocks of SOIprocessing techniques. Many of the components as seen in the blockdiagram exhibit large changes with temperature when processing a singlesignal. These errors are cancelled when the ratio of the two periods iscalculated. Such errors result, for example, from mismatch of gainresistors, changes in the value of the capacitor (e.g., capacitor 37)changes in the value of the current source 36, and changes in othercircuit components from device to device as well as the effectiveresistance of the long wire and the value of the supply voltage.

As indicated, the circuit can be implemented as a very small integratedcircuit chip with very few external connections. For a singletransducer, the chip requires only five pins, allowing use of a verysmall package. The functions of the external measuring circuit aresimpler than other implementations as the measuring circuit has tomeasure only two time periods which can be done with a comparator anddigital circuits with a much better accuracy and much less complexitythan any other measurement. The circuit can operate from a five voltsupply reducing the power dissipation compared with other circuitsrequiring much higher voltages. There is no need for voltage regulatorsor stable references as all of the pertinent features due to changes andso on are cancelled by performing a ratiometric indication.

It is well known based on the above, as to how the ratiometriccalculation can be performed as there are many circuits well known inthe art which are capable of providing division of, for example T_(P)divided by T_(R) as well as multiplication. All of this can beimplemented by a microprocessor or conventional circuits which arewidely available. As discussed herein, the ratiometric measurementaccording to an aspect of the present invention cancels outsubstantially all variations.

Referring to FIG. 5 there is shown a comparator 52, the positive inputor non-inverting comparator 52 is directed to resistor 51 where oneterminal of resistor 51 connected to the +5 volt supply which isequivalent to the VSM supply depicted in FIG. 1. Resistor 51 isequivalent to resistor 19 shown in FIG. 1. The positive input ofcomparator 52 is directed to wire 15 and hence to the output of thesensor interface as depicted in FIG. 2. The inverting input ofcomparator 52 is biased by connecting it to the common terminal of thevoltage divider consisting of resistors 53 and 54. The voltage dividerconsisting of resistors 53 and 54 supplies a voltage of 4.75 or 0.25volts below the supply voltage. This level corresponds to the middle ofthe pulses generated by the sensor interface. The pulses generated bythe sensor interface, are shown in FIG. 3B.

The output of the comparator is connected to an input of amicrocontroller or microprocessor 55 which includes a timer 56controlled by a crystal 57. The measuring system as indicated is locatedremotely from the sensor interface and as depicted in FIG. 1 isrepresented as measurement system 18. The measuring system is at thesurface of the ground if the interface 10 is placed in a drilledaperture or a drilled well associated with an oil well, for example. Inany event, the measuring circuit is at a normal temperature such as roomtemperature or ambient temperature. The measuring system determines thetwo periods in the pulses present in one complete cycle and thencalculates the ratio of these periods. This measurement can be done inmany ways, using typical time and frequency measurements. Two suchimplementations are depicted in the figures and are based on identifyingthe start and end of the cycle from the flat portion (for example nopulses, corresponding in duration to the sixty-four (64) pulse duration)of the waveform. This is shown in FIG. 3A. The sixty-four (64) referencelevel contains no pulses while the period indicative of the pressuretransducer output contains one thousand and twenty-four (1,024)measurement pulses, while the duration of the reference signal is ninehundred and sixty (960) reference pulses. As indicated above, the periodof the flat portion of the waveform is 204.8 milliseconds this is muchlonger than the periods of the measurement pulses which can vary between1.6 milliseconds to 6.4 milliseconds or the reference pulsesapproximately at a period of 3.2 milliseconds.

Still referring to FIG. 5, the signal from the interface which is thesignal shown in FIG. 3 is applied to the positive input of thecomparator 52, while the negative input of the comparator is biasedbelow the supply voltage at a level corresponding to the middle of thepulses generated by the sensor interface. This can be shown in FIG. 3B.The output of the comparator is applied to the interrupt input (INT) ofa microcontroller or a microprocessor 55. The microcontroller 55 timestamps each interrupt by reading the internal timer 56 as well as themicrocontroller counts these interrupts. By calculating the differencebetween the two successive interrupts, the microcontroller 55 identifiesfirst the flat portion of the waveform, as its duration is much longerthan any pulse in the sequence. After the flat portion is identified,the microcontroller 55 counts one thousand and twenty-four (1,024)pulses, determines their total duration and divides the result by onethousand and twenty-four (1,024), thus determining the period of thepulse as corresponding to the sensor data. Immediately after these onethousand and twenty-four (1,024) pulses, the microcontroller counts thenext nine hundred and sixty (960) pulses. It then determines their totalduration and divides the result by nine hundred and sixty (960) thusdetermining the period of the reference pulses.

The ratio of the two periods is then calculated to determine the valueof the quantity to be measured. This, for example, may be pressure inthe case of utilizing a pressure transducer or may be temperature in thecase of using a temperature transducer. Either the pressure or thetemperature transducer is arranged in a bridge circuit, as shown, forexample in FIG. 1. Therefore, one can utilize this technique to measurepressure, temperature or any other value which can be implemented as avoltage at the output of a bridge configuration.

Referring to FIG. 6 there is shown another exemplary method of measuringthe output of the sensor interface. Also seen in FIG. 6 there is aresistor 60 which is also equivalent to resistor 19 of FIG. 1. Theresistor 60 has one terminal coupled to the input of analog-to-digitalconverter 61 and the other terminal coupled to the +5 volt supply. Theinput of the analog-to-digital converter 61 is connected to wire 15 andthus connected to the sensor interface. The output of theanalog-to-digital converter 61 is connected to the input of amicrocontroller 62. The analog-to-digital converter 61 digitizes theincoming waveform. The sampling rate of the analog-to-digital converteris configured to be about ten (10) times that of the fastest pulses inthe sequence resulting in a sampling interval of one hundred and sixtymicroseconds (160 us). The flat portion of the waveform is easilyidentified as no major transitions occur through relatively longduration of 204.8 milliseconds.

Next the period of a complete cycle is determined as the time betweentwo successive flat portions. The waveform is then digitized for onecomplete cycle and the result stored. The computer then generates datafor a theoretical waveform, with the same structure as the real one.Thus, for example, the computer generates one thousand and twenty-four(1,024) measurement pulses, nine hundred and sixty (960) referencepulses and a flat portion corresponding to sixty-four (64) referencepulses. The periods of the measurement and the reference pulses arearbitrarily chosen. The cross correlation function of the two waveformsis then calculated while the two periods in the theoretical waveform arevaried until the cross correlation function shows a very short maximumvalue as a peak. The respective period values in the theoreticalwaveform corresponding to this peak represent the actual measurement andthe reference period. Thus, the reference period again is used toproduce the ratio between the pressure level period and the referenceperiod to produce an output indicative of pressure, while undesiredvariations are thereby cancelled.

The programming regarding the microcontroller shown in FIG. 5 and FIG. 6may be understood by one skilled in the art as the steps for producingand implementing the measuring system are clearly described. It isunderstood that there are other techniques which can be employed tomeasure the time period of both the frequency modulated pressuretransducer output and the frequency modulated reference signal leveloutput.

It should be understood by one skilled in the art that there are manyalterations, and variations of the above noted circuitry all of whichare deemed to be encompassed in the spirit and scope of the claimsappended hereto.

What is claimed is:
 1. A method, comprising: measuring a first monitoredcondition to generate a first monitored condition signal; converting thefirst monitored condition signal to a first frequency modulated signalhaving a first frequency; generating a second frequency modulated signalhaving a reference frequency; transmitting the first frequency modulatedsignal and the second frequency modulated signal using time divisionmultiplexing; and wherein a first ratio of the first frequency and thereference frequency is associated with the first monitored condition. 2.The method of claim 1, wherein transmitting the first frequencymodulated signal is performed during a first interval and transmittingthe second frequency modulated signal is performed during a secondinterval.
 3. The method of claim 2, wherein a third interval is disposedbetween the first interval and the second interval and the thirdinterval is used to differentiate between the first frequency modulatedsignal and the second frequency modulated signal.
 4. The method of claim1, further comprising: measuring a second monitored condition togenerate a second monitored condition signal; converting the secondmonitored condition signal to a third frequency modulated signal havinga second frequency; wherein transmitting the first frequency modulatedsignal and the second frequency modulated signal using the time divisionmultiplexing includes transmitting the third frequency modulated signalusing the time division multiplexing; and wherein a second ratio of thesecond frequency and the reference frequency is associated with thesecond monitored condition.
 5. The method of claim 4, whereintransmitting the third frequency modulated signal is performed during afourth interval.
 6. The method of claim 1, wherein transmitting thefirst frequency modulated signal and the second frequency modulatedsignal using time division multiplexing is performed over a transmissionmedium.
 7. The method of claim 6, wherein the transmission medium is awire.
 8. The method of claim 6, wherein the transmission medium is in arange of about one thousand meters to about ten thousand meters.
 9. Themethod of claim 1, wherein the first monitored condition is pressure.10. The method of claim 1, wherein the first monitored condition istemperature.
 11. A system, comprising: a first transducer configured to:receive a first monitored condition; measure the first monitoredcondition to generate a first monitored condition signal; and output thefirst monitored condition signal; a reference generator configured to:generate a reference signal; a converter operationally coupled to thefirst transducer and the reference generator, wherein the converter isconfigured to: convert the first monitored condition signal to a firstfrequency modulated signal having a first frequency; and convert thereference signal to a second frequency modulated signal having areference frequency; a transmitter operationally coupled to theconverter, wherein the transmitter is configured to: transmit the firstfrequency modulated signal and the second frequency modulated signalusing time division multiplexing; and wherein a first ratio of the firstfrequency and the reference frequency is associated with the firstmonitored condition.
 12. The system of claim 11, further comprising: amultiplexer coupled to the first transducer, the reference generator,the converter, and the transmitter, wherein the multiplexer isconfigured to: receive the first monitored condition signal; receive thereference signal; output the first monitored condition signal during afirst interval; and output the reference signal during a secondinterval.
 13. The system of claim 12, wherein the transmitter is furtherconfigured to: dispose a third interval between the first interval andthe second interval, wherein the third interval is used to differentiatebetween the first frequency modulated signal and the second frequencymodulated signal.
 14. The system of claim 11, further comprising: asecond transducer operationally coupled to the converter, wherein thesecond transducer is configured to: receive a second monitoredcondition; measure the second monitored condition to generate a secondmonitored condition signal; and output the second monitored conditionsignal; wherein the converter is further configured to: convert thesecond monitored condition signal to a third frequency modulated signalhaving a second frequency; wherein the transmitter is further configuredto: transmit the third frequency modulated signal using the timedivision multiplexing; and wherein a second ratio of the secondfrequency and the reference frequency is associated with the secondmonitored condition.
 15. The system of claim 14, further comprising: amultiplexer coupled to the first transducer, the second transducer, thereference generator, the converter, and the transmitter, wherein themultiplexer is configured to: receive the first monitored conditionsignal; receive the reference signal; receive the second monitoredcondition signal; output the first monitored condition signal during afirst interval; output the reference signal during a second interval;and output the second monitored condition signal during a thirdinterval.
 16. The system of claim 11, wherein the transmitter is furtherconfigured to: transmit the first frequency modulated signal and thesecond frequency modulated signal using the time division multiplexingover a transmission medium.
 17. The system of claim 16, wherein thetransmission medium is a wire.
 18. The system of claim 16, wherein thetransmission medium is in a range of about one thousand meters to aboutten thousand meters.
 19. The system of claim 11, wherein the firstmonitored condition is pressure.
 20. The system of claim 11, wherein thefirst monitored condition is temperature.